RF receiver with frequency tracking

ABSTRACT

A robust frequency drift tracking receiver. The received signal is translated to an intermediate frequency in the RF stage by a quadrature demodulator, and is then brought into the base band by a digital mixer made by a CORDIC. A base band processing stage allows for a synchronization of the receiver relative to the data frame, to estimate data and to output a counter-reaction signal to the CORDIC, obtained by integration of successive frequency corrections, with a predetermined step.

TECHNICAL DOMAIN

This invention relates to the domain of RF receivers and moreparticularly receivers for use in narrow band very long range and verylow throughput telecommunication systems such as those envisaged for theIoT (Internet of Things).

STATE OF PRIOR ART

In general, an RF receiver requires frequency tracking of the signaltransmitted by the transmitter. Central transmission and receptionfrequencies are never perfectly equal due to the imprecision ofoscillators and their temperature drift. Furthermore, the relativedisplacement velocity of the receiver relative to the transmittergenerates, a Doppler effect and therefore a frequency drift that has tobe compensated.

Different solutions have been put forward in prior art to estimate theCFO (Carrier Frequency Offset) between the transmitter and receiver tocompensate for it in the reception. In particular, it is known that afrequency tracking loop can be set up at the output from the quadraturedemodulation stage to control the frequency of VCO oscillators (VoltageControlled Oscillators) in this stage. Thus, the received signal iscontinuously returned into the base band or to a fixed intermediatefrequency. Different types of frequency tracking loops are described forexample in the article by F. D. Natali entitled “AFC trackingalgorithms” published in IEEE Trans. on Comm., Vol. COM-32, No. 8,August 1984.

More recently, application FR-A-2977943 disclosed a narrow band and lowthroughout transmission system in which a blind estimate of thefrequency offset is made from the RF signal itself after it has beenconverted to digital, and this frequency offset can be compensated onsamples of this signal.

However this frequency tracking loop requires relatively complexcalculations, which requires a large silicon surface area and leads tohigh consumption that is not easily compatible with the power levelavailable in a connected object.

Consequently, the purpose of this invention is to disclose an RFreceiver with a particularly simple and robust frequency tracking loop,suitable for low throughput transmission on a narrow band.

PRESENTATION OF THE INVENTION

This invention is defined by a frequency drift tracking receiverdesigned to receive packets of symbols modulating an RF signal, saidreceiver comprising an RF stage to translate the RF signal received atan intermediate frequency by means of a quadrature mixer and to digitisethe signal thus obtained, the receiver also comprising:

a digital mixer using a CORDIC to bring the frequency translated signalthus digitised into the base band, the CORDIC rotating the phase of eachsample as a function of an estimate of the intermediate frequency;

a base band digital processing module designed to be synchronised withpackets of symbols and to estimate data transmitted in said packets, andto determine the sign of the variation in the frequency drift for eachsymbol from the data thus estimated, the intermediate frequency beingestimated by integration of corrections of elementary frequency steps bya predetermined value modified by the signs thus determined.

Advantageously, the base band digital processing module comprises atleast three filters adapted to the shape of the pulse that was used tomodulate the RF signal, a first matched filter being centred on the zerofrequency, a second matched filter being offset relative to the firstmatched filter by a positive predetermined frequency difference and thethird matched filter being offset relative to the first matched filterby a negative predetermined frequency difference, the output signalsfrom the three matched filters being input firstly to a switch and to asynchronisation module controlling said switch to select a matchedfilter output signal.

Preferably, each symbol packet comprises a synchronisation preamble, apredetermined frame delimiter and a data frame, the synchronisationmodule searches for a frame delimiter in the output signals and selectsthe output signal in which the frame delimiter was found.

If the synchronisation module detects a frame delimiter in severaloutput signals, it selects the highest power output signal amongst theseoutput signals.

The selected output signal can then be resampled by a decimatorcontrolled by the synchronisation module, the decimator providingsamples at the symbol frequency.

Advantageously, the synchronisation module determines the highestamplitude sample in at least one sequence of samples corresponding to apulse, and controls the decimator so as to select this sample for eachsymbol.

If data were modulated by means of a DBPSK modulation, each sample atthe decimator output is multiplied with the conjugate of the previoussample by means of a DBPSK demodulator to provide the symbols of a BPSKmodulation constellation.

BPSK symbols at the output from the demodulator can be multiplied in anangular correction module by the conjugate of a magnitude characteristicof a rotation of the BPSK modulation constellation to output correctedsymbols.

In this case, an estimator can then make a hard estimate on thecorrected symbols to estimate the data.

The estimator can advantageously estimate data by determining the signof the real part of the corrected symbols.

The angular rotation estimating module advantageously multiplies BPSKsymbols at the output from the DBPSK demodulator with symbolscharacteristic of estimated data to supply a magnitude characteristic ofthe rotation of the BPSK constellation between two consecutive symbols.

The frequency drift tracking receiver can also comprise a moduledetermining the direction of variation of the frequency drift startingfrom the sign of the imaginary part of said characteristic magnitude.

If required, the characteristic magnitude can be filtered by a law passfilter before being input to said angular correction module.

The frequency drift tracking receiver can also comprise a second switchconnected to the synchronisation module and to the angular rotationestimating module that will output symbols forming the frame delimiterduring a synchronisation phase and said filtered characteristicmagnitude during reception of the data frame, to the angular correctionmodule.

The frequency pitch can advantageously be chosen to be less than 1/16Tin which T is the symbol period.

BRIEF DESCRIPTION OF THE DRAWINGS

Other characteristics and advantages of the invention will become clearafter reading a preferred embodiment of the invention with reference tothe attached figures among which:

FIG. 1 diagrammatically represents the architecture of an RF receiveraccording to one embodiment of the invention;

FIG. 2 diagrammatically represents the RF stage of the receiver in FIG.1;

FIG. 3 diagrammatically represents the base band digital processingstage of the receiver in FIG. 1;

FIG. 4 represents the format of a transmission packet;

FIG. 5 illustrates frequency tracking of the RF receiver during a framefor different examples of frequency drift;

FIG. 6 represents the packet error rate obtained at the output from thereceiver in FIG. 1, as a function of the signal to noise ratio fordifferent frequency drift examples.

DETAILED PRESENTATION OF PARTICULAR EMBODIMENTS

We will consider a receiver with the general architecture shown in FIG.1 for the remainder of this description.

The receiver comprises an RF stage 110, connected to the antenna 100,with the function of bringing the RF signal to a floating intermediatefrequency, f_(i) ^(a), and to sample it, a digital mixer 120, with thefunction of bringing the signal into the base band, and a base banddigital processing module 130 that will be described in detail later.

The digital mixer 120 is made from a CORDIC (COordinate Rotation DigitalComputer) that does a phase rotation at every instant to bring theintermediate frequency signal f_(i) ^(a) into the base band. Moreprecisely, if the sampling period is denoted T_(e) and the samplingindex is denoted k, the phase rotation made at instant kT_(e) is:Δφ=−2πf _(mix) ^(d) ·kT _(e) modulo 2π  (1)in which, f_(mix) ^(d)={circumflex over (f)}_(i) ^(a) is a frequencyvalue received by the CORDIC corresponding to an estimate of f_(i) ^(a).

The CORDIC achieves this phase rotation by elementary rotations ofvalues Δφ +dφ or −dφ (dφ being positive) depending on the sign of thephase rotation. The elementary phase increment dφ is chosen such thattan(dφ)=2^(−p) where p varies from 0 to N in which N is chosen to besufficiently large depending on the required degree of precision. Moreprecisely, in iteration p, the CORDIC receives an input vector v_(p) andcalculates an output vector v_(p+1) for each elementary rotation suchthat:v _(p+1) =R _(ϵδφ) v _(p)  (2)where R_(ϵδφ) is the rotation matrix of ϵ·dφ defined by:

$\begin{matrix}{R_{{ɛ\delta}\;\varphi} = \begin{pmatrix}1 & {{- ɛ}{.2}^{- p}} \\{ɛ{.2}^{- p}} & 1\end{pmatrix}} & (3)\end{matrix}$

It will be understood that the elementary rotation operation isparticularly easy because it is reduced to simple offsets andadditions/subtractions.

FIG. 2 diagrammatically represents the RF stage of FIG. 1;

This includes a loan noise amplifier LNA, 210, a quadrature mixer(mixers 221,222), translating the signal central frequency, f₀, to anintermediate frequency f_(i) ^(a)=f₀−f_(mix) ^(a) in which f_(mix) ^(a)is the frequency of the oscillator outputting sine curves in quadratureto mixers 221, 222. It is important to note that in this case the firstintermediate frequency f_(i) ^(a) is not a fixed frequency but that itvaries as a function of the chosen central frequency, the drift of thecentral frequency chosen for the transmission and the drift of theoscillator.

Signals in quadrature at the output from the mixer are then filtered bymeans of low pass filters 231, 232 before being amplified, and are thenconverted using, analogue-digital converters 251 and 252. If applicable,the signals in quadrature can then be passed through a low pass filterstep in digital and then a first decimation step (not shown).

In any case, the pairs of samples in quadrature, after being filteredand decimated when applicable, are input to the digital mixer 120 ofFIG. 1 in the form of a vector. Components in quadrature of the rotatedvector are then transmitted to the digital processing stage in base bandshown in FIG. 3.

For reasons of convenience, connections between the modules of thisfigure are indicated by simple arrows. However it will be understoodthat the processed samples are complex samples are consequently comprisea real part and an imaginary part.

The base band digital processing stage comprises an optionalfiltration/decimation stage 310, for example made in the form of a CIC(Cascaded Integrator Comb) filter. This CIC filter can eliminate anyinterferers and reduce the sampling rate.

The baseband digital signal is then filtered by three matched filters321, 322, 323 arranged in parallel. Filter 322 is a filter matched tothe shape of the transmitted filters, centred on the null frequency.Filters 321 and 323 are versions of the same matched filter, eachshifted by a frequency offset +ΔF, −ΔF from the null frequency. Ingeneral, it would be possible to allow for a plurality of matchedfilters with the same transfer function except, for a frequency offset,one of them being centred on the null frequency and the others beingcentred symmetrically about this frequency.

The matched filters are active simultaneously in a first phase calledthe synchronisation phase that will be described in detail later. At theend of this synchronisation phase, the matched filter that is bestcentred on the base band signal will be selected. This matched filterthen remains active throughout the remainder of the packet, the othermatched filters being deactivated or their outputs being inhibited.

In this case, output signals from the three matched filters 321-323 areinput firstly to the switch 330 and secondly to the synchronisationmodule 340.

The synchronisation module 340 determines the matched filter for whichthe output signal has the highest power during the synchronisationperiod, for example by comparing the energy of the different outputsignals during the duration of the synchronisation period. Thesynchronisation module 340 also uses the output signals to determine thebeginning of the data frame and selects the matched filter accordingly.Finally, the synchronisation module 340 determines symbol samplinginstants, each symbol giving rise to a plurality of samples at theoutput from the matched filters.

More precisely, FIG. 4 shows a transmission packet 400. This packetcomprises a synchronisation preamble, 410, a Start of Frame Delimiter(SFD) 420 and a data frame, 430. The synchronisation module determinesthe highest power output signal from the matched filters 321-323 duringthe synchronisation preamble (for example a sequence of alternatingbits). Knowing the sequence of symbols making up the SFD delimiter, thesynchronisation module attempts to identify the start of the data framein each output signal (for example using a correlation or a simplecomparison.

The synchronisation module then selects the matched filter for which theoutput signal was used to identify the SFD delimiter. If thesynchronisation module identifies the SFD delimiter in several outputsignals, the synchronisation module determines the one with the highestpower and selects the matched filter accordingly, using the switch 330.

Packet data are in the form of DBPSK (Differential Binary Phase ShiftKeying) symbols or possibly BPSK (Binary Phase Shift Keying) symbols,each symbol modulating a pulse filtered by a Pulse Shaping Filter (PSF).

At the receiver end, at the output from the selected matched filter,each symbol gives rise to a plurality K of successive samples in which Kis the ratio between the sample rate at the output from module 310 andthe symbol rate, namely

$K = {\frac{T}{T_{ɛ}}.}$The synchronisation module 340 determines the optimum sampling instantamong the plurality of successive instants (that with the highestamplitude).

Signal samples at the output from the selected matched filter areresampled by the decimator 350. To achieve this, the synchronisationmodule 340 supplies the optimum instant to a decimator 350, withdecimation factor K. Consequently, output signals from the decimator 350are at the symbol rate.

The modulation used by the transmitter may be a BPSK modulation, orpreferably a DBPSK modulation.

When the transmitter uses a differential modulation (DBPSK), the samplesoutput from the decimator 350 are firstly subjected to a differentialdemodulation at 355. This is done in a manner known in itself bycalculating the Hermitian product of the current sample and the previoussample. Obviously, the differential demodulator 355 is not present inthe case of a BPSK type direct modulation.

Output samples from the decimator 350, possibly after differentialdemodulation 355, are BPSK signals. An angular correction is made onthem at 360, to compensate for the rotation of the modulationconstellation as described below. The symbols are then estimated bymaking a hard decision using the estimator 370, from the samples thuscorrected.

The module 380 estimates a magnitude characteristic of the angularrotation of the modulation constellation starting from estimated symbolsand output samples from the decimator 350. This characteristic magnitudeis filtered by means of a low pass filter (LPF) before being input tothe angular correction module 360.

Said characteristic magnitude is also input to the module 390 thatdeduces the direction of variation of the frequency drift, ϵ_(n),between two consecutive symbols.

The integrator module 395 summates successive frequency corrections, thesuccessive corrections being equal to ϵ_(n)·δf in which δf is apredetermined frequency step. This sum of successive corrections isinput to the digital mixer 120 as an estimate of the intermediatefrequency,

.

Operation of the base band digital processing stage will be describedfor the case of a DBPSK modulation. The signal transmitted by theemitter can then be expressed in the following form:

$\begin{matrix}{{s_{Ts}(t)} = {A\;{\cos\left( {{2\pi\; f_{0}t} + \alpha} \right)}{\sum\limits_{k}{d_{k}{p_{0}\left( {t - {kT}} \right)}}}}} & (4)\end{matrix}$in which A is the amplitude, of the transmitted signal, f₀ is thecentral frequency of the signal, α the phase at the origin, p₀(t) thepulse shape (for example in Root Raised Cosine—RRC), T is the symbolperiod and d_(k) are the DBPSK symbols. Remember that DBPSK symbols areobtained from data bits b_(k) using:b _(k) ′=b _(k) ⊕b _(k−1)′d _(k)=+1 if b _(k)′=0d _(k)=−1 if b _(k)′=1  (5)and conversely:d _(k) d _(k−1)=+1

b _(k)=0d _(k) d _(k−1)=−1

b _(k)=1  (5′)The shape of the resampled signal at the output from the decimator 350is then as follows:

$\begin{matrix}{{s_{dbd}(n)} = {{B\;{{\exp\left( {i\left( {{2{\pi\left( {f_{0} - f_{1}} \right)}{nT}} + \varphi} \right)} \right)} \cdot {\sum\limits_{k}{d_{n}{p_{1}\left( {\left( {n - k} \right)T} \right)}}}}} + {N(n)}}} & (6)\end{matrix}$in which B is the amplitude of the signal at the output from theselected matched filter, f₁ is the sum of the frequency of the analoguemixer (f_(mix) ^(a)) and the frequency of the digital mixer (f_(mix)^(d)), φ is a phase dependent on the phase of the carrier and phases ofthe mixers, p₁(t) is the self-correlation of the pulse shape p₀(t) (orequivalently, the signal p₀(t) filtered by the matched filter), and N(n)is a noise sample.

After differential demodulation, the samples output from module 355 areexpressed in the following form:

$\begin{matrix}{\sigma_{n} = {{{s_{dbd}(n)}{{s{^\circ}}_{dbd}\left( {n - 1} \right)}} = {{B^{2}{\exp\left( {i\; 2\;{\pi\left( {f_{0} - f_{1}} \right)}T} \right)}d_{n}d_{n - 1}} + {B\;{\exp\left( {{i\; 2\;{\pi\left( {f_{0} - f_{1}} \right)}{nT}} + {i\;\varphi}} \right)}{{N{^\circ}}\left( {n - 1} \right)}} + {B\;{\exp\left( {{{- i}\; 2\;{\pi\left( {f_{0} - f_{1}} \right)}\left( {n - 1} \right)T} - {i\;\varphi}} \right)}{N(n)}} + {{N(n)}.{{N{^\circ}}\left( {n - 1} \right)}}}}} & (7)\end{matrix}$

If it is assumed that the signal to noise ratio is sufficient, in otherwords the terms in which the noise appears can be neglected, then:σ_(n) =B ²exp(i2π(f ₀ −f ₁)T)d _(n) d _(n−1)  (8)

When frequency tracking is done by the digital mixer, we obtain:|2π(f ₀ −f ₁)T|<<π/2  (9)

If, for the moment, we ignore the angular correction in module 360, theestimator 370 estimates the BPSK values, c_(n), using the hard decision:

=sgn(Re(σ_(n)))  (10)the data bits being deduced traditionally

$= \frac{1 -}{2}$with the modulation convention defined in (5).

The module 380 estimates the instantaneous angular rotation of themodulation constellation starting from:

=arg(σ_(n)

)  (11-1)or more precisely, estimates the corresponding characteristic magnitude:a _(n) =B ²

=σ_(n)

  (11-2)

It is important to note that multiplying σ_(n) by the estimated symbols

make the process independent of the influence of data.

The complex material a_(n) is advantageously filtered by a low passfilter (LPF), for example a recursive filter with a forgetting factor.The complex magnitude thus filtered

is used to compensate for rotation of the constellation in the module360 by calculating the Hermitian product:σ_(n) ^(c)=σ_(n)

  (12)

Thus when angular compensation is active, the corrected samples, σ_(n)^(c), in other words samples corrected by the angular correction, areused in the expression (10).

The module 390 determines the direction of variation (or the frequencydrift) between consecutive samples:ϵ_(n)=sgn(Im(σ_(n)

))  (13)

This sign calculation is particularly simple, it makes it possible torobustly monitor the frequency variation. The frequency correction ismade in steps of δf, in which:

$\begin{matrix}{{\delta\; f} = {\frac{\delta\;\varphi}{2\;\pi}\frac{1}{T}}} & (14)\end{matrix}$in which δφ is a predetermined phase skip.

Preferably, δf< 1/16T will be chosen such that the corresponding phaseskip,

${{{\delta\varphi}} \leq \frac{\pi}{8}},$does not disturb the bit estimate.

The integrator module 395 then calculates the frequency f_(mix) ^(d),sum of the supposed initial frequency f_(mix) ^(d) ^(_) ^(init) andsuccessive corrections:

$\begin{matrix}{f_{mix}^{d} = {f_{mix}^{d\_ init} + {\delta\; f{\sum\limits_{n}ɛ_{N}}}}} & (15)\end{matrix}$

This frequency is input to the digital mixer 120 as an estimate of theintermediate frequency,

. The digital mixer rotates the phase Δφ_(n+1) obtained by recurrence:Δφ_(n+1)=Δφ_(n)−2πf _(mix) ^(d) T  (16)

Using this frequency tracking, the signal output from the digital mixerat frequency Δf=f₀−f₁, where f₁=f_(mix) ^(a)+f_(mix) ^(d), is keptwithin the spectral response of the selected matched filter.

The angular rotation of the modulation constellation is compensatedfirstly during the synchronisation phase and secondly during receptionof data.

During the synchronisation phase, the receiver knows the sequence ofpilot symbols c_(n) ^(p) of the preamble. The sequence of output symbolsfrom each matched filter is correlated with the sequence of pilotsymbols. Successive correlation peaks can be used to determinedecimation instants at the symbol frequency in the decimator 350.

Furthermore, knowledge of pilot symbols can be used to estimate theangular rotation using:a _(n) ^(p)=σ_(n)(c _(n) ^(p))*  (17)

The symbols c_(n) ^(p) are output directly from the synchronisation 340to the angular correction module 360. Thus, during the synchronisationphase, retroaction of the output from the estimator 370 to the angularcorrection module is prevented. During this phase, there is no moreretroaction from the integrator module 395 to the CORDIC 120.

The switch 365 changes position between the synchronisation phase andthe data reception phase. More precisely, during the synchronisationphase, it transmits values of correlation peaks (possible filtered usinga low pass filter) output from the synchronisation module 340 and(during the data reception phase), symbols

output from the tow pass filter 385, to the angular correction module360.

The angular rotation is compensated using the Hermitian product σ_(n)^(c)=σ_(n)(a_(n) ^(p))* during the synchronisation phase and using theσ_(n) ^(c)=σ_(n)

product during the data reception phase. This compensation corrects theprecise rotation of the constellation due to the offset between thefrequency Δf and the real frequency difference Δf_(ext), represented bythe magnitude a_(n) ^(p) (during the synchronisation phase) and

(during the data reception phase).

The receiver disclosed above is designed to receive DBPSK symbols.However, the man skilled in the art will understand that an embodimentin which BPSK symbols are received could be envisaged as an alternative.In this case, as mentioned above, the differential demodulation module355 is eliminated and the magnitude a_(n) calculated by the module 380is obtained by a_(n)=σ_(n)

, in which

in this case is the BPSK symbol corresponding to bit

estimated by the estimator 370 (in other words

=+1 if

=1 and

=−1 if

=0). As above, the modulation effect due to data is thus neutralised.

A numeric example is given below illustrating an application of theinvention to the domain of the “internet of things”. The signal istransmitted in the ISM band at 868 MHz. The central frequency of thesignal is around 869.5 MHz in a 48 kHz band. The format of transmissionpackets is shown in FIG. 4. The throughput is 100 bits/s, in other wordsT=10 ms and the modulation is a DBPSK modulation. The mix frequency ofthe analogue mixer is 868.6 MHz and therefore the intermediate frequencyif there is no frequency tracking is of the order of 900 000 Hz.Analogue-digital analogue converters 241-242 of the RF stage outputsamples at a sampling frequency of 13.572 MHz. A first decimation step,is done in the RF stage and a second decimation step is done in thefiltering/decimation module 310. At the output, the samples are outputat a rate of 600 Hz, namely 6 samples per symbol. In this case, thematched filters are RRC filters. The central filter 322 is centred on 0Hz, the filter is centred on +50 Hz and the filter 323 is centred on −50Hz. The decimator 350 resamples with a factor of 6, to return to onesample per symbol. The frequency step δf is chosen to the equal to 1 Hz,in other words the frequency cannot, vary by more than ±1 Hz at eachsymbol.

In the following, it will be assumed that the receiver configurationparameters are the same as above.

FIG. 5 illustrates frequency tracking of the RF receiver during a framefor different examples of frequency drift;

Example 510 corresponds to a zero frequency offset at the beginning ofthe packet and zero drift during the packet.

Example 520 corresponds to a zero frequency offset at the beginning ofthe packet and 20 Hz/s drift during the packet.

Example 530 corresponds to a 20 Hz/s frequency offset at the beginningof the packet and 20 Hz/s drift during the packet.

Example 540 corresponds to a 50 Hz/s frequency offset at the beginningof the packet and 20 Hz/s drift during the packet.

It can be seen that the intermediate frequency controlling the digitalmixer (CORDIC) starts to follow the frequency drift after the end of thesynchronisation period (40 symbols). The frequency correction by meansof the CORDIC is not active during the synchronisation period.

It can be seen in examples 510-530, that the matched filter centred on 0Hz is selected, the drift being caught up by the digital mixer laterwhen the initial offset is larger (see 530 compared with 520).

In example 540, the matched filter centred on +50 Hz is selected. Onceagain, the intermediate frequency starts to follow the frequency driftat the end of the synchronisation phase.

FIG. 6 represents the performances of the receiver in FIG. 1 in terms ofthe PER (Packet Error Rate) as a function of the signal to noise ratio,for different examples of frequency drift.

It will be seen that the Packet Error Rate (PER) remains less than 10%when the signal to noise ratio is more than 10 dB, even in the case of alarge frequency drift.

The invention claimed is:
 1. A frequency drift tracking receiverdesigned to receive packets of symbols modulating an RF signal, saidreceiver comprising an RF stage to translate the RF signal received atan intermediate frequency with a quadrature mixer and to digitise thesignal thus obtained, said frequency drift tracking receiver comprising:a digital mixer using a Coordinate Rotation Digital Computer (CORDIC) tobring the frequency translated signal thus digitised into a base band,the CORDIC making a phase rotation of each sample as a function of anestimate of the intermediate frequency; and a base band digitalprocessing module, implemented by circuitry, designed to be synchronisedwith packets of symbols and to estimate data transmitted in saidpackets, and to determine a sign of a variation in frequency drift foreach symbol from the data thus estimated, the intermediate frequencybeing estimated by integration of corrections of elementary frequencysteps by a predetermined value modified by the signs thus determined. 2.The frequency drift tracking receiver according to claim 1, wherein thebase band digital processing module comprises at least three filtersmatched to a shape of a pulse that was used to modulate the RF signal, afirst matched filter being centred on a zero frequency, a second matchedfilter being offset relative to the first matched filter by a positivepredetermined frequency difference and a third matched filter beingoffset relative to the first matched filter by a negative predeterminedfrequency difference, output signals from the three matched filtersbeing input firstly to a switch and to a synchronisation module,implemented by the circuitry, controlling said switch to select amatched filter output signal.
 3. The frequency drift tracking receiveraccording to claim 2, wherein each symbol packet comprises asynchronisation preamble, a predetermined frame delimiter and a dataframe, the synchronisation module searches for a frame delimiter inmatched filter output signals and selects the output signal in which theframe delimiter was found.
 4. The frequency drift tracking receiveraccording to claim 3, wherein the selected matched filter output signalis then resampled by a decimator, which is implemented by the circuitryand controlled by the synchronisation module, the decimator providingsamples at a symbol frequency.
 5. The frequency drift tracking receiveraccording to claim 4, wherein the synchronisation module determines ahighest amplitude sample in at least one sequence of samplescorresponding to the pulse, and controls the decimator so as to selectthis sample for each symbol.
 6. The frequency drift tracking receiveraccording to claim 5, wherein data were modulated by a DifferentialBinary Phase Shift Keying (DBPSK) modulation, each sample at a decimatoroutput is multiplied with a conjugate of the previous sample by a DBPSKdemodulator to provide symbols of a Binary Phase Shift Keying (BPSK)modulation constellation.
 7. The frequency drift tracking receiveraccording to claim 6, wherein BPSK symbols at the output from thedemodulator are multiplied in an angular correction module, which isimplemented by the circuitry, by a conjugate of a magnitudecharacteristic of a rotation of the BPSK modulation constellation tooutput corrected symbols.
 8. The frequency drift tracking receiveraccording to claim 7, wherein an estimator, implemented by thecircuitry, makes a hard estimate on the corrected symbols to estimatedata.
 9. The frequency drift tracking receiver according to claim 8,wherein the estimator estimates data by determining a sign of a realpart of the corrected symbols.
 10. The frequency drift tracking receiveraccording to claim 9, wherein the characteristic magnitude is filteredby a low pass filter before being input to said angular correctionmodule.
 11. The frequency drift tracking receiver according to claim 10,further comprising a second switch, connected to the synchronisationmodule and to an angular rotation estimating module that outputs symbolsforming the frame delimiter during a synchronisation phase and saidfiltered characteristic magnitude during reception of the data frame, tothe angular correction module.
 12. The frequency drift tracking receiveraccording to claim 8, wherein an angular rotation estimating module,implemented by the circuitry, multiplies BPSK symbols at the output fromthe DBPSK demodulator with symbols characteristic of estimated data tosupply the magnitude characteristic of the rotation of the BPSKconstellation between two consecutive symbols.
 13. The frequency drifttracking receiver according to claim 12, further comprising a module,implemented by the circuitry, determining a direction of variation ofthe frequency drift starting from a sign of an imaginary part of saidmagnitude characteristic.
 14. The frequency drift tracking receiveraccording to claim 2, wherein the synchronisation module detects a framedelimiter in several output signals, and selects a highest power outputsignal amongst these output signals.
 15. The frequency drift trackingreceiver according to claim 1, wherein a frequency pitch is less than inwhich is a symbol period.